Power supply apparatus

ABSTRACT

In a power supply apparatus for performing constant current driving of a light emitting diode which is a load circuit, a constant current circuit is disposed on a path for driving the load circuit. A charge pump circuit which is a voltage generating circuit outputs a driving voltage to the light emitting diode. A monitoring circuit monitors the voltage across the two ends of the constant current circuit. This monitoring circuit includes a voltage source which generates a threshold voltage that follows the fluctuation of the voltage at which the constant current circuit can operate stably, compares the voltage across the two ends of the constant current circuit and the threshold voltage generated by the voltage source, and outputs a comparison result Vs to a control unit. The control unit controls the charge pump circuit on the basis of the output of the monitoring circuit.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation application of the U.S. patentapplication Ser. No. 11/792,267, filed Jun. 4, 2007, the contents ofwhich are incorporated by reference herein in their entirety, andpriority to which is claimed under 35 U.S.C. §120; further, applicationSer. No. 11/792,267 is a U.S. national stage of application No.PCT/JP2005/021241, filed on Nov. 18, 2005 priority to which claimedherein under 35 U.S.C. §119 (a) and 35 U.S.C. §365(b) and priority towhich is also claimed from Japanese Application No. 2004-350871, filedDec. 3, 2004; and Japanese Application No. 2005-143522, filed May 17,2005; the disclosures of which are also incorporated herein byreference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a power supply apparatus.

2. Description of the Related Art

In small information terminals such as a mobile phone and a PDA(Personal Digital Assistance) of recent years, there are devices thatrequire a voltage higher than the output voltage of the battery such asa light emitting diode (hereinafter also referred to as a LED) used as aback light of a liquid crystal, for example. In these small informationterminals, a Li ion battery is often used, and the output voltagethereof is typically about 3.5 V, and is about 4.2 V in the fullycharged state. However, the LED requires a voltage higher than thebattery voltage as the driving voltage thereof. In this manner, when avoltage higher than the battery voltage is required, the battery voltageis boosted with use of a power supply apparatus of booster type using acharge pump circuit or the like, thereby to obtain a voltage needed fordriving a load circuit such as a LED.

In driving the LED with such a power supply apparatus, the control ofthe luminescence brightness thereof is stabilized by connecting aconstant current circuit on a path for driving the LED and keeping theelectric current flowing through the LED to be constant (See PatentDocument 1).

In order to achieve stable operation of the constant current circuitconnected to the cathode terminal of the LED, the transistorconstituting the constant current circuit need to operate in a constantcurrent region. Here, the constant current region of a transistor refersto an active region in the case of a bipolar transistor, and refers to asaturation region in the case of a field effect transistor (hereinafterreferred to as FET). The transistors constituting the constant currentcircuit is disposed in series between the cathode terminal of the LEDand the grounded terminal. In order that this transistor may operate inthe constant current region, the cathode terminal of the LED must bekept at a predetermined voltage or higher. Hereinafter, the voltage atwhich the constant current circuit can operate stably will be simplyreferred to as a stable operation voltage.

Here, in a power supply apparatus for driving the LED, the case of usinga charge pump circuit capable of switching the boosting ratio isconsidered (See Patent Document 2). When the battery voltage that isinput into the charge pump circuit lowers, the output voltage of thecharge pump circuit, namely, the voltage at the anode terminal of theLED, also lowers. In accordance therewith, the voltage at the cathodeterminal of the LED, which has been lowered by the forward voltage Vffrom the voltage at the anode terminal of the LED, also lowers, so thatthe constant current circuit cannot be operated stably. Therefore, inthis case, the constant current circuit can be stably operated bymonitoring the voltage at the cathode terminal of the LED and switchingthe boosting ratio of the charge pump circuit so that the voltage at thecathode terminal of the LED may not become lower than the predeterminedstable operation voltage.

-   [Patent Document 1] Japanese Patent Application Laid-Open No.    2004-22929-   [Patent Document 2] Japanese Patent Application Laid-Open No.    H6-78527

In the case of switching the boosting ratio of the charge pump circuitby monitoring the voltage at the cathode terminal of the LED asdescribed above, a threshold voltage corresponding to the stableoperation voltage of the constant current circuit must be set, and thevoltage at the cathode terminal must be controlled to be higher thanthis threshold voltage.

However, since the element characteristics of the transistors and theresistors constituting the constant current circuit fluctuate due tovariation of the semiconductor manufacturing processes and thetemperature, the stable operation voltage of the constant currentcircuit also fluctuates in accordance therewith. For this reason, thethreshold voltage must be set to be higher in consideration of theprocess variations and the temperature characteristics. For example,when the stable operation voltage of the constant current circuitfluctuates within a range of ±0.1 V at the maximum with the designedvalue of 0.3 V at the center, the voltage of 0.4 V or higher is set asthe threshold voltage so as to ensure the margin.

Here, in the case in which this threshold voltage is set to be 0.4 V,when the stable operation voltage of the constant current circuitbecomes 0.2 V due to the process variations, the voltage of the cathodeterminal, which should be stabilized inherently at 0.2 V or higher, isstabilized to be 0.4 V or higher. In other words, though the boostingratio should inherently be raised when the voltage of the cathodeterminal becomes 0.2 V or lower, the boosting ratio must nevertheless beraised at the time the voltage of the cathode terminal becomes 0.4 V orlower.

Since the efficiency of the charge pump circuit becomes deterioratedaccording as the boosting ratio becomes higher, the efficiency of thewhole circuit will be deteriorated when a margin is set at the thresholdvoltage for achieving stable operation of the constant current circuitas described above.

SUMMARY OF THE INVENTION

The present invention has been made in view of these problems, and ageneral purpose thereof is to provide a power supply apparatus which iscapable of higher-efficiency operation by appropriately setting thedriving voltage of a load circuit.

One embodiment of the present invention relates to a power supplyapparatus. This power supply apparatus is a power supply apparatus forperforming constant current driving of a load circuit and includes aconstant current circuit which is disposed on a path for driving theload circuit; a voltage generating circuit which outputs a drivingvoltage to the load circuit; a monitoring circuit which monitors avoltage across the two ends of the constant current circuit; and acontrol unit which controls the driving voltage that is output from thevoltage generating circuit. The monitoring circuit includes a thresholdvoltage source which generates a threshold voltage that follows thefluctuation of the voltage at which the constant current circuit canoperate stably, and outputs to the control unit a result obtained bycomparing the voltage across the two ends of the constant currentcircuit and the threshold voltage generated by the threshold voltagesource. The control unit controls the voltage generating circuit on thebasis of the output of the monitoring circuit.

According to this embodiment, when the element characteristics of thetransistors and the resistors constituting the constant current circuitfluctuate due to variation of the semiconductor manufacturing processesor temperature fluctuation, and the stable operation voltage of theconstant current circuit fluctuates in accordance therewith, a suitabledriving voltage can be output by controlling the voltage generatingcircuit on the basis of the threshold voltage that follows thefluctuation.

The constant current circuit may include a current output terminal towhich the load circuit to be driven is connected; an operation amplifierhaving a first input terminal to which a predetermined reference voltageis applied; a first transistor having a control terminal to which anoutput voltage of the operation amplifier is applied and having one endconnected to the current output terminal; a first resistor connected tothe other end of the first transistor and having one end to which apredetermined fixed voltage is applied; and a feedback path which feedsthe electric potential of the connection point of the first transistorand the first resistor back to a second input terminal of the operationamplifier. The threshold voltage source may include a constant currentsource which outputs a predetermined constant current; a secondtransistor which is disposed in series on a path of the constant currentthat is output from the constant current source; and a second resistorhaving one end to which the fixed voltage is applied and having theother end to which the second transistor is connected, and may outputthe voltage of the connection point of the second transistor and theconstant current source as the threshold voltage.

By allowing the threshold voltage source which generates the thresholdvoltage and the principal part of the constant current circuit to havesimilar constructions, when the characteristics of the first resistorand the first transistor of the constant current circuit fluctuate, thethreshold voltage which is output from the threshold voltage sourcechanges in accordance with this fluctuation, so that thevoltage-generating circuit can be appropriately controlled.

The second transistor and the first transistor, and the second resistorand the first resistor may be formed respectively by pairing on asemiconductor integrated circuit.

In the voltage source which generates the threshold voltage and in theconstant current circuit, when the elements corresponding to each otherare formed respectively by pairing, the characteristics fluctuation ofthe corresponding elements can be matched, thereby enabling generationof a more appropriate threshold voltage.

The constant current that is output from the constant current source maybe set within a range at which the second transistor operates in aconstant current region.

The monitoring circuit may include a voltage comparator which comparesthe voltage across the two ends of the constant current circuit and thethreshold voltage generated by the voltage source, and an offset voltageadjusting circuit which adjusts the offset voltage of the voltagecomparator. In this case, by adjusting the offset voltage of the voltagecomparator, an error voltage between the stable operation voltage of theconstant current circuit and the threshold voltage can be cancelled.

The constant current circuit may further include a reference resistordisposed on a path of a reference current which accords with theconstant current that is output from the constant current source andhaving one end to which the fixed voltage is applied, and the voltageappearing in the other end of the reference resistor may be applied tothe first input terminal of the operation amplifier as the referencevoltage. In this case, when the constant current that is output from theconstant current source fluctuates, the electric current which isgenerated by the constant current circuit and the threshold voltagefluctuate simultaneously, so that the error can be cancelled.

The offset voltage adjusting circuit may adjust a differential currentof the voltage comparator. By increasing or decreasing the differentialcurrent of the voltage comparator, the offset voltage can be shifted inboth directions of the positive direction and the negative direction, sothat the threshold voltage can be adjusted more accurately.

Also, the offset voltage adjusting circuit may include a main currentsource which generates a tail current to be supplied to a differentialpair of the voltage comparator; a first variable current source whichgenerates a first variable current and changes one of the differentialcurrents generated by the differential pair; and a second variablecurrent source which generates a second variable current and changes theother of the differential currents generated by the differential pair.

The main current source, and the first and second variable currentsources of the offset voltage adjusting circuit may be integrallyconstructed to include a second constant current source and a currentmirror circuit which duplicates the constant current generated by thisconstant current source with an adjustable mirror ratio, and theelectric current duplicated by the current mirror may be output to thevoltage comparator as the tail current, the first variable current, andthe second variable current.

In this case, when the mirror ratio of the current mirror circuit isadjusted by trimming of the interconnection and the fuse, the ratios ofthe first and second variable currents to the tail current can bechanged, so that the offset voltage of the voltage comparator can besuitably adjusted. Further, since the three electric currents aregenerated on the basis of one constant current, the relative variationcan be restrained to be small.

The offset voltage adjusting circuit may include a plurality ofadjusting transistors disposed in parallel to the transistorsconstituting the differential pair of the voltage comparator andtrimmable fuses disposed on the respective electric current paths of theadjusting transistors. Also, the offset voltage adjusting circuit mayinclude a plurality of adjusting transistors disposed in parallel to thetransistors constituting the current mirror load connected to thedifferential pair of the voltage comparator and trimmable fuses disposedon the respective electric current paths of the adjusting transistors.

Fine adjustment of the transistor sizes of the differential pair and thecurrent mirror load with use of the adjusting transistors can adjust theoffset voltage of the voltage comparator.

The voltage generating circuit may be a charge pump circuit which switchamong a plurality of boosting ratios, and the control unit may switch aboosting ratio of the charge pump circuit on the basis of a result ofvoltage comparison of the monitoring circuit. Also, the voltagegenerating circuit may be a switching regulator circuit, and the controlunit may control a switching operation of the switching regulatorcircuit so that the voltage across the two ends of the constant currentcircuit and the threshold voltage will be equal to each other in themonitoring circuit.

The load circuit may be a light emitting diode; and the constant currentcircuit may be connected to a cathode terminal of the light emittingdiode; and the monitoring circuit may monitor the voltage at the cathodeterminal of the light emitting diode. In this case, the constant currentdriving of the light emitting diode can be efficiently performed.

Another embodiment of the present invention is a light emittingapparatus. This light emitting apparatus includes a light emitting diodeand the above-described power supply apparatus for performing constantcurrent driving of the light emitting diode. According to thisembodiment, the constant current driving of the light emitting diode canbe efficiently performed, so that the lifetime of the battery can beextended.

It is to be noted that any arbitrary combination or rearrangement of theabove-described structural components and so forth is effective as andencompassed by the present embodiments.

Moreover, this summary of the invention does not necessarily describeall necessary features so that the invention may also be asub-combination of these described features.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments will now be described, by way of example only, withreference to the accompanying drawings which are meant to be exemplary,not limiting, and wherein like elements are numbered alike in severalFigures, in which:

FIG. 1 is a view showing an overall construction of a power supplyapparatus and a light emitting apparatus according to an embodiment.

FIG. 2 is a circuit diagram showing a construction of the charge pumpcircuit of FIG. 1.

FIG. 3 is a circuit diagram showing a construction of the constantcurrent circuit and the monitoring circuit of FIG. 1.

FIG. 4 is a view showing the current voltage characteristics of the FETwhich is the first transistor.

FIG. 5 is a view showing a relationship between the battery voltagewhich will be the input voltage of the charge pump circuit and theefficiency η.

FIG. 6 is a circuit diagram showing a construction of the monitoringcircuit and the constant current circuit.

FIG. 7 is a circuit diagram showing a construction of a voltagecomparator and an offset voltage adjusting circuit.

FIG. 8 is a circuit diagram showing a modified example of the voltagecomparator and the offset voltage adjusting circuit of FIG. 7.

FIG. 9 is a circuit diagram showing a construction of a voltagecomparator whose offset voltage can be adjusted.

FIG. 10 is a view showing a modified example of the power supplyapparatus of FIG. 1.

FIG. 11 is a circuit diagram showing apart of the construction of thedriving circuit of FIG. 10.

FIG. 12 is a view showing a modified example of the constant currentcircuit of FIG. 3.

DETAILED DESCRIPTION OF THE INVENTION

The invention will now be described based on preferred embodiments whichdo not intend to limit the scope of the present invention but exemplifythe invention. All of the features and the combinations thereofdescribed in the embodiment are not necessarily essential to theinvention.

FIG. 1 shows a light emitting apparatus 1000 according to an embodimentof the present invention. In the subsequent drawings, the sameconstituent elements will be denoted with the same numerals, and thedescription will not be repeated appropriately. The light emittingapparatus 1000 is mounted, for example, on an electronic equipment suchas a mobile phone terminal or a PDA, and functions as a back light of aliquid crystal. FIG. 1 illustrates a liquid crystal panel 510 togetherwith the light emitting apparatus 1000. A light emitting diode 300 isplaced on the back surface of the liquid crystal panel 510, andfunctions as the back light.

This light emitting apparatus 1000 includes the light emitting diode 300which is a light emitting element, and a power supply apparatus 100 fordriving the light emitting diode 300. The light emitting apparatus 1000is mounted on an information terminal that is driven by a battery 500,and the power supply apparatus 100 boosts the battery voltage Vbat thatis output from the battery 500, so as to generate a driving voltage Voutthat is needed in driving the light emitting diode.

The power supply apparatus 100 includes, as input and output terminals,an input terminal 102 into which the battery voltage Vbat is input, anoutput terminal 104 that is connected to the anode terminal of the lightemitting diode 300 and outputs the output voltage Vout obtained byboosting the battery voltage Vbat, and a LED terminal 106 that isconnected to the cathode terminal of the light emitting diode 300.

The power supply apparatus 100 includes a charge pump circuit 10 and adriving circuit 20 thereof. The charge pump circuit 10 boosts thebattery voltage Vbat that is input from the input terminal 102 at apredetermined boosting ratio, and generates the output voltage Vout fromthe output terminal 104. This charge pump circuit 10 is constructed tobe capable of switching among a plurality of boosting ratios. In thepresent embodiment, it is assumed that the boosting ratio can beswitched among the three ways of 1, 1.5, and 2.

FIG. 2 is a circuit diagram showing a construction example of the chargepump circuit 10. The charge pump circuit 10 includes a first capacitorC1, a second capacitor C2, and a first switch SW1 to a ninth switch SW9for controlling the connection state of these capacitors. Hereinafter,these switches will be generally called switches SW unless there is aneed to distinguish these switches specifically. The first capacitor C1and the second capacitor C2 are set to have equal (capacitance values,and are externally attached to the outside of an integrated circuit.

The first switch SW1 to the ninth switch SW9 can be constructed withfield effect transistors FET of N-type or P-type, and can be operated asswitching elements by controlling the conduction state between the drainand the source with a voltage applied to the gate. In this charge pumpcircuit 10, the state of on/off of the first switch SW1 to the ninthswitch SW9 is switched by a controlling signal Vcnt that is output froma driving circuit 20. Here, it is assumed that the controlling signalVcnt is input into each of the first switch SW1 to the ninth switch SW9,though not illustrated in FIG. 2.

The charge pump circuit 10 is constructed so as to be capable ofswitching among a plurality of boosting ratios as described above. Here,the operation of the charge pump circuit 10 in accordance with theboosting ratios will be described.

When the boosting ratio is set to be 1, the first switch SW1, the thirdswitch SW3, the seventh switch SW7, and the eighth switch SW8 arestationary turned on by the driving signal Vcnt that is output from thedriving circuit 20, and all the other switches are turned off. As aresult of this, the input terminal 102 and the output terminal 104 arebrought into a conduction state by the switches that are turned on, sothat the battery voltage Vbat that is applied to the input terminal 102is output from the output terminal 104, whereby the boosting ratio willbe set to be 1.

Therefore, the controlling signal Vcnt that is output from the drivingcircuit 20 will not be a switching signal that repeats on and off withlapse of time but will be a constant voltage when the boosting ratio isset to be 1.

Next, the operation when the boosting ratio is set to be 1.5 will bedescribed. In the case in which the boosting ratio is larger than 1,namely, in the case of performing a boosting operation, the charge pumpcircuit 10 repeats a first period and a second period having differentconnection states of the switches.

In the first period, the first capacitor C1 and the second capacitor C2are connected in series and are charged with the battery voltage Vbat byturning the first switch SW1, the fifth switch SW5, and the sixth switchSW6 on and turning all the other switches off. Since the capacitancevalues of the first capacitor C1 and the second capacitor C2 are equal,each of the two capacitors will be charged with Vbat/2 which is half ofthe battery voltage Vbat.

In the second period, the second switch SW2 and the seventh switch SW7,the fourth switch SW4 and the eighth switch SW8 are turned on, and allthe other switches are turned off. At this time, the first capacitor C1and the second capacitor C2 are connected in parallel between the inputterminal 102 and the output terminal 104. As a result of this, a sum ofthe battery voltage Vbat applied to the input terminal 102 and thevoltage of charging the capacitors will be output from the outputterminal 104. In the first period, since the first capacitor C1 and thesecond capacitor C2 are charged with the voltage Vbat/2, a voltage ofVbat+Vbat/2=1.5×Vbat will be eventually output from the output terminal104.

In this manner, the charge pump circuit 10 magnifies the battery voltageVbat by 1.5 times for output by repeating the first period and thesecond period.

Next, the operation when the boosting ratio is set to be 2 will bedescribed.

In the first period, the first switch SW1 and the ninth switch SW9, thethird switch SW3 and the sixth switch SW6 are turned on, and all theother switches are turned off. The first capacitor C1 and the secondcapacitor C2 are connected in parallel between the input terminal 102and the grounded terminal GND, whereby each is charged with the batteryvoltage Vbat.

In the second period, the second switch SW2 and the seventh switch SW7,the fourth switch SW4 and the eighth switch SW8 are turned on, and allthe other switches are turned off. As a result of this, the firstcapacitor C1 and the second capacitor C2 are connected in parallelbetween the input terminal 102 and the output terminal 104. A sum of thebattery voltage Vbat applied to the input terminal 102 and the voltageof charging the capacitors will be output from the output terminal 104.In the first period, since the first capacitor C1 and the secondcapacitor C2 are each charged with the battery voltage Vbat, a voltageof Vbat+Vbat=2×Vbat will be output from the output terminal 104.

In this manner, the charge pump circuit 10 magnifies the battery voltageVbat by 2 times for output by repeating the first period and the secondperiod.

Returning to FIG. 1, the driving circuit 20 sets the boosting ratio ofthe charge pump circuit 10, and controls the boosting operation, namely,the connection state of the switches SW of the charge pump circuit 10.This driving circuit 20 includes a constant current circuit 22, acontrol unit 24, a first oscillator 26, a second oscillator 28, and amonitoring circuit 30.

The constant current circuit 22 is connected to the cathode terminal ofthe light emitting diode 300 via a LED terminal 106. Since theluminescence brightness of the light emitting diode 300 is determined bythe electric current Iled that flows through the light emitting diode300, the constant current circuit 22 controls the electric current Iledso that the luminescence brightness of the light emitting diode 300 willbe a desired value.

The monitoring circuit 30 monitors the voltage across the two ends ofthe constant current circuit 22 in order to switch the boosting ratio ofthe charge pump circuit 10. The monitoring circuit 30 compares thevoltage across the two ends of the constant current circuit 22 and apredetermined threshold voltage, and outputs the result of comparison tothe control unit 24. In the present embodiment, the voltage across thetwo ends of the constant current circuit 22 corresponds to the voltagebetween the grounded terminal and the LED terminal 106. Though thedetails will be described later, the control unit 24 switches theboosting ratio of the charge pump circuit 10 on the basis of the outputVs from the monitoring circuit 30.

FIG. 3 is a circuit diagram showing a construction of the constantcurrent circuit 22 and the monitoring circuit 30.

The constant current circuit 22 includes a first transistor M1, a firstresistor R1, and an operation amplifier 40. The first transistor M1 is aMOSFET (Metal Oxide Semiconductor Field Effect Transistor) of N-type.

A predetermined reference voltage Ve is applied to the non-invertinginput terminal of the operation amplifier 40. This reference voltage Veis a voltage for controlling the luminescence brightness of the lightemitting diode 300. Regarding the first transistor M1, the outputvoltage of the operation amplifier 40 is applied to the gate which is acontrol terminal, and the drain is connected to the LED terminal 106.One end of the first resistor R1 is connected to the source of the firsttransistor M1, and a predetermined ground voltage is applied to theother end thereof. The electric potential Vr1 at the connection point ofthe first transistor M1 and the first resistor R1 is fed back to theinverting input terminal of the operation amplifier 40.

The voltage Vr1 applied to the first resistor R1 is fed back to theinverting input terminal of the operation amplifier 40, and the feedbackis applied so that the voltages at the inverting input terminal and atthe non-inverting input terminal will be equal to each other. Thereforethe voltage applied to the first resistor R1 will be approximated to thereference voltage Ve. When the voltage Vr1 applied to the first resistorR1 is equal to the reference voltage Ve, an electric current Idrv=Ve/R1flows through the first resistor R1. This electric current Idrv is noneother than the electric current Iled that flows through the lightemitting diode 300 via the first transistor M1 and the LED terminal 106.

In this manner, the constant current circuit 22 generates a constantcurrent Iled=Ve/R1 based on the reference voltage Ve, and controls theelectric current. Iled that flows through the light emitting diode 300.

Here, in order that this constant current circuit 22 may generate anelectric current in a stable manner, the first transistor M1 must beoperated in a constant current region. The constant current region meansthe saturation region when the transistor is a field effect transistorFET, and means an active region when the transistor is a bipolartransistor.

When the voltage Vled of the LED terminal 106 lowers, the electricpotential difference between the two ends of the first transistor M1,namely the drain-source voltage, will be small, whereby the firsttransistor M1 will operate in a non-saturation region. In thenon-saturation region, the electric current that flows between the drainand the source will be dependent on the drain-source voltage, so thatthe constant current circuit 22 will not operate as a constant currentcircuit. Therefore the luminescence brightness of the light emittingdiode 300 cannot be stabilized.

For this reason, the monitoring circuit 30 monitors so that the voltageVled of the LED terminal 106 may not become lower than a predeterminedthreshold voltage Vth, as shown in FIG. 3. This threshold voltage Vth isset within a range such that the first transistor M1 operates in aconstant current region (saturation region), namely within a range suchthat the constant current circuit 22 can generate a predeterminedconstant electric current.

The monitoring circuit 30 includes a voltage comparator 50 and athreshold voltage source 52 that outputs the threshold voltage Vth. Thevoltage Vled of the LED terminal 106 and the threshold voltage Vth areinput into the voltage comparator 50, whereby a high level is outputwhen Vled>Vth holds, and a low level is output when Vled<Vth holds. Theoutput Vs of this voltage comparator 50 is input into the control unit24.

The control unit 24 raises the boosting ratio of the charge pump circuit10 by one step when a state in which the voltage Vs output from themonitoring circuit 30 is at the low level, namely a state in whichVled<Vth holds, continues for a predetermined period of time. In otherwords, when the voltage Vs output from the monitoring circuit 30 becomesthe low level while the operation is carried out at a boosting ratio of1, the control unit 24 sets the boosting ratio to be 1.5. Similarly,when the voltage Vs output from the monitoring circuit 30 becomes thelow level while the operation is carried out at boosting ratio of 1.5,the control unit 24 sets the boosting ratio to be 2.

As a result of this, even in the case in which the battery voltage Vbatlowers due to the discharging of the battery 500 and the voltage Vled atthe cathode terminal of the light emitting diode 300 lowers inaccordance therewith, the boosting ratio can be appropriately switched.When the boosting ratio is set to be high, the output voltage Vout thatis output from the output terminal 104 rises, so that the voltage Vledat the LED terminal 106 can be made higher than the threshold voltageVth, whereby the constant current circuit 22 can be stably operated.

The threshold voltage Vth that is output from the threshold voltagesource 52 is set to be a stable operation voltage of the constantcurrent circuit 22, namely, within a range such that the firsttransistor M1 operates in a constant current region (saturation region).For example, this threshold voltage Vth is set to be 0.3 V.

Here, the element characteristics and the circuit characteristics of thefirst transistor M1, the first resistor R1, and the operation amplifier40 constituting the constant current circuit 22 fluctuate in accordancewith the variation of the semiconductor manufacturing process or thetemperature. FIG. 4 is a view showing the current voltagecharacteristics (IV characteristics) of the FET which is the firsttransistor M1, where the longitudinal axis represents the drain-sourcecurrent Ids, and the lateral axis represents the drain-source voltageVds.

In the drawing, with an average current voltage characteristics IVm1, itis a saturation region when the drain-source voltage is higher than thevoltage Vx1, and it is a non-saturation region when the drain-sourcevoltage is lower than the voltage Vx1. Now, assuming that the currentvoltage characteristics change to the current voltage characteristicsIVm2 due to the variation of the semiconductor manufacturing process orthe temperature change, the boundary voltage of the saturation regionand the non-saturation region will also be shifted to Vx2 in accordancetherewith.

The voltage across the two ends of the constant current circuit 22 willbe a sum of the voltage drop Vr1 by the first resistor R1 and thedrain-source voltage of the first transistor M1. Therefore, inaccordance with the fluctuation of the current voltage characteristicsof the first transistor M1, the stable operation voltage of the constantcurrent circuit 22 will also change. Similarly, this stable operationvoltage will also change in accordance with the variation of theresistance value of the first resistor R1.

Assuming that the current voltage characteristics of the firsttransistor M1 fluctuate between IVm1 and IVm2 of FIG. 4 due to thevariation of the semiconductor manufacturing process or the temperaturechange, the voltage for stably operating the constant current circuit 22will fluctuate within a range from Vth1=Ic×R1+Vx1 to Vth2=Ic×R1+Vx2.

In the case in which the threshold voltage Vth generated by thethreshold voltage source 52 of the monitoring circuit 30 is a constantvalue, the threshold voltage Vth must be set to be Vth1=Ic×R1+Vx1 inconsideration of the margin in order to operate the constant currentcircuit 22 stably in all of the ranges in which the current voltagecharacteristics of the first transistor M1 fluctuate.

Here, the efficiency of the charge pump circuit 10 will be studied. FIG.5 is a view showing the battery voltage Vbat which will be the inputvoltage of the charge pump circuit 10 and the efficiency η.

Here, a case in which the threshold voltage Vth generated by thethreshold voltage source 52 is fixed to a certain voltage Vth1 will beconsidered. When the boosting ratio is 1, the relationship between thebattery voltage Vbat and the voltage Vled at the LED terminal 106 isrepresented by Vbat=Vled+Vf using the forward voltage Vf of the lightemitting diode 300. Now, when the relationship Vbat<Vbat1 (=Vth1+Vf) issatisfied in accordance with the decrease in the battery voltage Vbat,the voltage Vled at the LED terminal 106 will be <Vth1, so that theboosting ratio will be switched from 1 to 1.5.

In this manner, when the threshold voltage Vth is fixed to a certainvoltage Vth1, even in a case in which the characteristics of the firsttransistor M1 vary and the stable operation voltage of the constantcurrent circuit 22 becomes lower than the certain voltage Vth1, theboosting ratio will be switched to 1.5 in a state in which Vbat<Vbat1holds, thereby leaving room for improvement in view of the efficiency.

Therefore, in order to improve the efficiency of the charge pump circuit10, the threshold voltage source 52 of the monitoring circuit 30according to the present embodiment is constructed to generate athreshold voltage Vth that follows the fluctuation in thecharacteristics of the first transistor M1 and the first resistor R1.

Returning to FIG. 3, the threshold voltage source 52 includes a secondtransistor M2, a second resistor M2, and an electric current source 54.

The second transistor M2, the second resistor R2, and the electriccurrent source 54 are connected in series, and a constant voltage Icgenerated by the electric current source 54 is allowed to flow throughthe second transistor M2 and the second resistor R2. The power supplyvoltage Vdd is applied to the gate of the second transistor M2.

This threshold voltage source 52 outputs the voltage at the connectionpoint of the second transistor M2 and the electric current source 54 asthe threshold voltage Vth. The drain-source voltage Vds2 of the secondtransistor M2 is determined by the constant current Ic, and the voltageVr2 appearing in the second resistor R2 is given by Vr2=Ic×R2. As aresult of this, the threshold voltage Vth can be represented asVth=Ic×R2+Vds2.

In this manner, in the threshold voltage source 52, the construction ofthe principal part for generating the threshold voltage Vth isapproximately the same as that of the constant current circuit 22. On asemiconductor integrated circuit, the first resistor R1 and the secondresistor R2 are preferably formed to be close to each other by beingpaired. Similarly, the first transistor M1 and the second transistor M2are preferably formed to be close to each other so as to be paired.

In this manner, by forming the principal constructions of the constantcurrent circuit 22 and the threshold voltage source 52 to be the sameand forming the resistors and the transistors constituting the circuitsby pairing, the amount of fluctuation of the characteristics of thecorresponding elements can be made to be approximately equal to eachother.

As a result of this, in the case in which the current voltagecharacteristics of the first transistor M1 fluctuate and the boundaryvoltage Vx between the saturation region and the non-saturation regionof the first transistor M1 fluctuates, the boundary voltage Vx betweenthe saturation region and the non-saturation region of the secondtransistor M2 will also fluctuate, so that the threshold voltage Vth canbe changed to follow the fluctuation in the characteristics of the firsttransistor M1.

Similarly, in the case in which the resistance value of the firstresistor R1 fluctuates due to the variation of the semiconductormanufacturing process or the temperature change, the resistance value ofthe second resistor R2 will also fluctuate in a similar manner, so thatthe threshold voltage Vth will also follow the fluctuation in thecharacteristics of the second resistor R2.

By constructing the monitoring circuit 30 as described above, even ifthe stable operation voltage of the constant current circuit 22fluctuates due to the fluctuation of the element characteristics causedby the process variation or the temperature change, the thresholdvoltage Vth is generated in accordance with the fluctuation, so that anoptimum setting of the boosting ratio can be made in the control unit24.

As a result of this, it means that the voltage for switching theboosting ratio can be appropriately set within a range from Vbat1 toVbat2 as shown in FIG. 5, so that the efficiency of the charge pumpcircuit 10 can be improved. Similarly, the switching of the boostingratio from 1.5 to 2 is performed with an optimum voltage, so that theefficiency can be improved.

Next, a technique of performing a further optimum setting of theboosting ratio with use of the monitoring circuit 30 will be described.FIG. 6 is a circuit diagram showing a construction of the monitoringcircuit 30 and the constant current circuit 22. The monitoring circuit30 of FIG. 6 is provided with an offset voltage adjusting circuit 56.

The voltage comparator 50 compares the voltage Vled at the LED terminal106 which is the voltage across the two ends of the constant currentcircuit 22 and the threshold voltage Vth generated by the thresholdvoltage source 52. The offset voltage adjusting circuit 56 adjusts theoffset voltage ΔV of the voltage comparator 50.

The constant current source 58, the transistor M3, and the transistor M5of FIG. 6 correspond to the electric current source 54 of FIG. 3. Theconstant current source 58 generates a reference current Iref. Thetransistors M3, M5 constitute a current mirror circuit, and a constantcurrent Ic proportional to the reference current Iref is output to thethreshold voltage source 52. Also, since the transistors M3, M4, M5constitute a current mirror circuit, the reference current Iref′ that isinput into the constant current circuit 22 is an electric current whichcorresponds to the constant current Ic that is output to the thresholdvoltage source 52. In the constant current circuit 22, the referencevoltage Ve that is applied to the non-inverting input terminal of theoperation amplifier 40 is generated by allowing the reference currentIref′ to flow through the reference resistor Rref. Namely, the referencevoltage Ve is given by Ve=Iref′×Rref, and the driving current Idrv willbe Idrv=Iref′×Rref/R1 and is proportional to the reference currentIref′. The reference resistor Rref is also preferably formed by beingpaired with the first resistor R1 and the second resistor R2.

When the value of the reference current Iref′ fluctuates due to theprocess variation or the temperature change and the value of the drivingcurrent Idrv fluctuates, the saturation drain-source voltage of thefirst transistor M1, namely the stable operation voltage of the constantcurrent circuit 22, fluctuates. In the monitoring circuit 30 of FIG. 6,the threshold voltage Vth generated by the threshold voltage source 52and the driving current Idrv generated by the constant current circuit22 are both set on the basis of the reference current Iref generated bythe constant current source 58. Therefore, when the reference currentIref fluctuates and the stable operation voltage of the constant currentcircuit 22 fluctuates, the threshold voltage Vth will also fluctuate inaccordance therewith. At this time, the voltage that is input into thevoltage comparator 50 will be shifted in the same direction, so that theprocess variation can be cancelled.

FIG. 7 is a circuit diagram showing a construction of the voltagecomparator 50 and the offset voltage adjusting circuit 56. The voltagecomparator 50 includes transistors M30 to M36, constant current sources80, 90, 92, and an amplifying stage 86. The transistors M30, M31constitute an input differential pair, and the respective gates 82, 84correspond to the two input terminals of the voltage comparator 50. Thedrains of the transistors M30, M31 are connected to the current mirrorload including the transistors M33, M34 provided as a constant-currentload. The transistors M33, M34 are a current mirror circuit in which thegates and the sources thereof are connected in common to those of thetransistors M32, and a constant electric current generated by theconstant current source 80 flows through each transistor.

The drains of the transistors M33, M34 are connected respectively to thesources of the transistors M35, M36. The gates of the transistors M35,M36 are connected in common, and the gate and the drain of thetransistor M35 are connected. Constant-current sources 90, 92 areconnected respectively to the drains of the transistors M35, M36. Thedrain of the transistor M36 is connected to an amplifying stage 86. Thedrain current of the transistor M36 will be a differential currentobtained by differential amplification of the gate voltages of thetransistors M30, M31. The amplifying stage 86 amplifies the differencebetween the electric current generated by the constant current source 92and the drain current of the transistor M36, and outputs it from theoutput terminal 44 of the voltage comparator 50. Here, the constructionof the voltage comparator 50 shown in FIG. 7 is one example, so thatvoltage comparators of various other circuit forms can be used.

The offset voltage adjusting circuit 56 adjusts the offset voltage ΔV byadjusting the differential current of the voltage comparator 50. Theoffset voltage adjusting circuit 56 includes transistors M20 to M25,fuses Rf1 to Rf4, and a constant current source 94.

The transistor M21 functions as a main current source which generates atail current Iss which will be supplied to the input differential pair(transistors M30, M31) of the voltage comparator 50. Also, thetransistors M22, M23 and the fuses Rf1, Rf2 generate a first variablecurrent Iv1 and functions as a first variable current source thatincreases one differential current Id1 that is generated by thedifferential pair (M30, M31). Also, the transistors M24, M25 and thefuses Rf3, Rf4 generate a second variable current Iv2 and functions as asecond variable current source that increases one differential currentId2 that is generated by the differential pair (M30, M31).

The constant current source 94 generates a constant electric currentIc1. The transistors M20 to M25 constitute a current mirror circuit inwhich the gates and the sources are connected in common, and duplicatesthe constant current Ic1 in accordance with a mirror ratio thatcorresponds to the size ratio of each transistor to generate the tailcurrent Iss, the first variable current Iv1, and the second variablecurrent Iv2. The electric current value of the first variable currentIv1 is made variable by a cut state of the fuses Rf1, Rf2. The electriccurrent value of the second variable current Iv2 also changes similarlyin accordance with a cut state of the fuses Rf3, Rf4. For example, whenthe size ratio of the transistors M21, M22, M23 is set to be 100:2:1,the first variable current Iv1 can be adjusted within a range of 3%, 2%,1%, and 0% relative to the tail current Iss. The same applies to thetransistors M24, M25 as well.

The first variable current Iv1 is supplied to one transistor M30 side ofthe input differential pair of the voltage comparator 50, and the secondvariable current Iv2 is supplied to the other transistor M31 side of thedifferential pair. According to the offset voltage adjusting circuit 56of FIG. 7, the differential current of the voltage comparator 50 can beadjusted by the cut state of the fuses Rf1 to Rf4. When the differentialcurrent of the operation amplifier is adjusted, the voltage-currentcharacteristics of the input differential pair will be shifted, so thatthe offset voltage ΔV can be adjusted.

Returning to FIG. 6, generally the resistance value and the transistorcharacteristics of a semiconductor integrated circuit vary in accordancewith a semiconductor manufacturing process, and the magnitude of thisvariation is varied in accordance with the layout of each element, thekind of the semiconductor manufacturing process, and others. For thisreason, the mirror ratio of the transistors M4, M5 that constitute thecurrent mirror circuit also has a variation dependent on thesemiconductor manufacturing process.

For example, it is assumed that the values of the reference currentIref′ and the constant current Ic vary relatively within a range ofabout ±2%. This variation will be an error of the threshold voltage Vthgenerated by the threshold voltage source 52 and the stable operationvoltage of the constant current circuit 22. When the threshold voltageVth deviates from the optimum value, it is not desirable because it willinvite deterioration in the efficiency of the power supply apparatus 100as described above.

Therefore, by trimming the fuses Rf1 to Rf4 in the offset voltageadjusting circuit 56 shown in FIGS. 6 and 7, the offset voltage ΔV ofthe voltage comparator 50 is shifted in the direction of canceling thevariation of the mirror ratio of the transistors M4, M5. As describedabove, the first variable current Iv1 and the second variable currentIv2 can be adjusted within a range of 0%, 1%, 2%, and 3% relative to thetail current Iss of the differential pair. Also, the sign (positive ornegative) of the offset voltage ΔV will be reversed between the case inwhich the fuses Rf1, Rf2 are trimmed and the case in which the fusesRf3, Rf4 are trimmed.

By changing the offset voltage ΔV of the voltage comparator 50, thefluctuation of the stable operation voltage of the constant currentcircuit 22 caused by the process variation can be cancelled, whereby thedeterioration in the efficiency of the power supply apparatus 100 can berestrained.

In other words, in the offset voltage adjusting circuit 56, the sizeratio of the transistors M21 to M25 that generate the tail current Iss,the first variable current Iv1, and the second variable current Iv2 maybe set so as to be capable of sufficiently reducing the error betweenthe threshold voltage Vth and the voltage needed for stable operation ofthe constant current circuit 22.

FIG. 8 is a circuit diagram showing a modified example of the voltagecomparator 50 and the offset voltage adjusting circuit 56 of FIG. 7.

In FIG. 7, the first variable current Iv1 is supplied to one transistorM30 side of the differential pair, and the second variable current Iv2is supplied to the other transistor M31 side of the differential pair.In contrast, in FIG. 8, the first variable current Iv1 is supplied toone transistor M35 side of the current mirror load (M35, M36) connectedto the differential pair (M30, M31), and the second variable current Iv2is supplied to the other transistor M36 side of the current mirror load.In the case in which the differential current is adjusted in thismanner, the offset voltage ΔV of the voltage comparator 50 can beadjusted as well, and an effect similar to that of FIG. 7 can beobtained. The same applies to the case in which the first variablecurrent Iv1 and the second variable current Iv2 are supplied to otherpositions capable of adjusting the differential current.

The offset voltage ΔV of the voltage comparator 50 can also be adjustedby the circuit shown in FIG. 9. FIG. 9 is a circuit diagram showing aconstruction of the voltage comparator 50 whose offset voltage can beadjusted. This voltage comparator 50 is constructed integrally with theoffset voltage adjusting circuit 56. In the present embodiment, theoffset voltage adjusting circuit includes transistors M40 to M43 foradjustment and fuses Rf1 to Rf4.

The transistors M40 to M43 for adjustment are disposed in parallel withthe transistors M30, M31 constituting the differential pair of thevoltage comparator 50. Trimmable fuses Rf1 to Rf4 are disposed on anelectric current path of the transistors M40 to M43 which aretransistors for adjustment.

According to the voltage comparator 50 of FIG. 9, the size of thetransistors constituting the differential pair of the voltage comparator50 can be changed substantially by the trimming state of the fuses Rf1to Rf4. As a result of this, the differential current is adjusted, andthe offset voltage ΔV can be shifted.

Also, though the size of the transistors of the differential pair isadjusted in FIG. 9, the size of the transistors M33, M34 constitutingthe current mirror load may be adjusted as a modified example. Namely, aplurality of transistors for adjustment are disposed in parallel withthe transistors M33, M34, and a trimmable fuse is disposed on anelectric current path of each transistor of these transistors foradjustment. By constructing an offset voltage adjusting circuit 56 withthe transistors for adjustment and the fuses and trimming the fuses, thedifferential current can be adjusted, and the offset voltage ΔV can beadjusted.

Similarly, the size of the transistors M35, M36 may be adjusted in placeof the transistors M33, M34.

Returning to FIG. 1, the control unit 24 sets the boosting ratio of thecharge pump circuit 10, and generates a controlling signal Vcnt inaccordance with the set boosting ratio. This control unit 24 monitorsthe output signal Vs of the monitoring circuit 30, and raises theboosting ratio when a state in which the output signal Vs is at a lowlevel continues for a predetermined period of time. In the presentembodiment, the control unit 24 raises the boosting ratio of the chargepump circuit 10 by one step when the output signal Vs of the monitoringcircuit 30 continues to be at a low level for 2 ms.

The periodic signal needed for the control unit 24 to generate thecontrolling signal Vcnt and to perform the measurement of time is outputfrom a first oscillator 26 and a second oscillator 28. The firstoscillator 26 and the second oscillator 28 are each provided with anenabling terminal not illustrated in the drawings, and are constructedto be capable of stopping the operation.

When the control unit 24 performs the boosting operation with the chargepump circuit 10, namely, when the boosting ratio is set to be 1.5 or 2,the controlling signal Vcnt will be a switching signal that turns thefirst switch SW1 to the ninth switch SW9 on and off. The firstoscillator 26 generates a first periodic signal Vosc1 having a frequencyneeded for this switching signal. For example, the frequency of thisfirst periodic signal Vosc1 is set to be 1 MHz.

Also, the control unit 24 generates a second periodic signal Vosc2having a frequency needed for measuring the time 2 ms in monitoring theoutput signal Vs of the monitoring circuit 30. Since the time of about 2ms can be measured with use of a frequency of about several ten kHz, itis assumed in the present embodiment that this second periodic signalVosc2 is set to have a frequency of 64 kHz.

The driving circuit 20 uses either one of the first oscillator 26 andthe second oscillator 28 by switching in accordance with the boostingratio of the charge pump circuit 10. For this reason, the control unit24 outputs an enabling signal that controls the on-off on the enablingterminals of the first oscillator 26 and the second oscillator 28 inaccordance with the boosting ratio of the charge pump circuit 10.

Hereinafter, the operation in switching the boosting ratio of the chargepump circuit 10 in the driving circuit 20 will be described.

When the battery voltage Vbat output from the battery 500 issufficiently high, the boosting ratio is set to be 1. Now, when thebattery voltage Vbat lowers due to electric power consumption, thevoltage Vled of the LED terminal 106 also lowers. In the monitoringcircuit 30, the threshold voltage Vth output from the threshold voltagesource 52 and the voltage Vled of the LED terminal 106 are compared and,when Vled<Vth holds, the monitoring circuit 30 outputs a low level as anoutput signal Vs.

When the boosting ratio of the charge pump circuit 10 is set to be 1, itis sufficient to stationarily turn the first switch SW1, the thirdswitch SW3, the seventh switch SW7, and the eighth switch SW8 on in thecharge pump circuit 10, so that the first periodic signal Vosc1 having afrequency of 1 MHz is not needed. For this reason, when the boostingratio is 1, the control unit 24 turns the first oscillator 26 off andallows only the second oscillator 28 to operate, thereby to performmeasurement of time by using the second periodic signal Vosc2.

When the output signal Vs of the monitoring circuit 30 continues to beat the low level for 2 ms, the control unit 24 switches the boostingratio to 1.5. When the boosting ratio is higher than 1, a switchingsignal that repeats on and off must be generated as a controlling signalVcnt to be output to the charge pump circuit 10 as described above. Atthis time, the control unit 24 needs the first periodic signal Vosc1,and hence turns the first oscillator 26 on. When the boosting ratio is1.5, the control unit 24 performs measurement of time for monitoring thestate of the output signal Vs of the monitoring circuit 30 by using thefirst periodic signal Vosc1. At this time, since the second periodicsignal Vosc2 is not needed, the control unit 24 turns the secondoscillator 28 off.

Further, also when the battery voltage Vbat lowers and the boostingratio is set to be 2, the control unit 24 turns only the firstoscillator 26 on, and performs generation of the controlling signal Vcntand measurement of time of 2 ms on the basis of the first periodicsignal Vosc1.

The consumption current of an oscillator is dependent on the frequency.The higher the frequency is, the larger the consumption current will be.In other words, the consumption current of the first oscillator 26 islarger than the consumption current of the second oscillator 28. Forthis reason, with the driving circuit 20 according to the presentembodiment, in the case of performing the boosting operation, the firstoscillator 26 that oscillates at 1 MHz is turned on, so as to generatethe controlling signal Vcnt and to perform measurement of time forsetting the boosting ratio. On the other hand, when the boosting ratiois 1, there is no need to generate a signal having a high frequency asthe controlling signal Vcnt, so that, by switching to the secondoscillator 28 having a smaller consumption current, the consumptioncurrent of the circuit can be reduced, thereby achieving a higherefficiency.

As shown above, the construction and the operation of the power supplyapparatus 100 according to the present embodiment have been described.With the power supply apparatus 100 according to the present embodiment,even if the stable operation voltage of the constant current circuit 22fluctuates due to fluctuation of the element characteristics caused byprocess variation or temperature change, the threshold voltage Vth isgenerated in accordance with the fluctuation, so that an optimumboosting ratio can be set in the control unit 24 and an appropriateoutput voltage Vout can be generated.

The above-described embodiment is an exemplification, and it will beunderstood by those skilled in the art that various modified examplescan be made in the combination of the constituent elements and thetreating processes thereof, and that those modified examples are alsowithin the scope of the present invention.

FIG. 10 is a view showing a modified example of the power supplyapparatus 100. In the power supply apparatus 100 of FIG. 10, a switchingregulator 70 is used instead of the charge pump circuit 10 as a voltagegenerating circuit. This switching regulator 70 is a voltage generatingcircuit that performs energy conversion between an inductor and acapacitor by on and off of a switching element, so as to raise the inputvoltage.

From the control unit 24, a switching signal Vpwm subjected to pulsewidth modulation (hereinafter referred to as PWM) is output, and theon-off of the switching element of the switching regulator 70 iscontrolled by this switching signal, whereby the output voltage Vout isstabilized to a desired voltage value.

FIG. 11 is a circuit diagram showing a part of the construction of thedriving circuit 20 FIG. 10. The construction of the constant currentcircuit 22 and the threshold voltage source 52 is the same as that ofFIG. 3. In the driving circuit 20 of FIG. 11, the voltage Vled of theLED terminal 106 and the threshold voltage Vth output from the thresholdvoltage source 52 are input into an error amplifier 60. The erroramplifier 60 amplifies the error between the voltage Vled and thethreshold value Vth, and outputs it to the control unit 24 as an errorvoltage Verr.

On the basis of this error voltage Verr, the control unit 24 generates aswitching signal Vpwm. The control unit 24 includes a voltage comparator62, a driver 64, and an oscillator 66. The oscillator 66 outputs aperiodic signal Vsaw having a triangular wave shape, and the voltagecomparator 62 generates a signal that has been subjected to pulse widthmodulation by comparing this periodic signal Vsaw and the error voltageVerr. On the basis of the output of the voltage comparator 62, thedriver 64 generates a switching signal for driving the switchingregulator 70.

As a result of this, the output voltage Vout of the switching regulator70 is adjusted so that the voltage Vled of the LED terminal 106 will beapproximated to the threshold voltage Vth. Therefore, an unnecessarilyhigh output voltage Vout is not generated while the constant currentcircuit 22 is being stably operated, so that a highly efficientoperation can be realized.

FIG. 12 is a view showing a modified example of the constant currentcircuit 22. In the constant current circuit 22 of FIG. 12, the drivingcurrent Idrv is generated with the power supply voltage Vdd being afixed voltage. The constant current circuit 22 of FIG. 12 includes atransistor M1, a first resistor R1, a reference resistor Rref, and anoperation amplifier 40. A LED 300 to be driven is connected to the LEDterminal 106. The reference resistor Rref is disposed on a path of thereference current Iref, and a predetermined fixed voltage Vdd is appliedto one end of the reference resistor Rref. The voltage Ve appearing inthe other end of the reference resistor Rref is applied to thenon-inverting input terminal of the operation amplifier 40. Thetransistor M1 has a gate to which the output voltage of the operationamplifier 40 is applied and a source that is connected to the LEDterminal 106. The first resistor R1 is connected to the source of thetransistor M1, and the fixed voltage Vdd is applied to one end of thefirst resistor R1. The electric potential at the connection pointbetween the transistor M1 and the first resistor R1 is fed back to theinverting input terminal of the operation amplifier 40. Even in the casein which the constant current circuit 22 is constructed in this manner,a driving current Idrv given by Idrv=Iref×Rref/R1 can be generated.

In the present embodiment, all of the elements constituting the powersupply apparatus 100 and the light emitting apparatus 1000 may beintegrated, or alternatively a part thereof may be constructed withdiscrete components, or a plurality of components may be made into amodule as one package. Which part should be integrated may be determinedaccording to the costs, the occupied area, and the like.

In the embodiment, description has been made on a case in which the LEDis driven by the power supply apparatus 100. However, the load circuitis not limited to this, so that other light emitting elements such as anorganic EL may be driven, and further a current-driven device other thana light emitting element can also be driven.

In the present embodiment, the transistor to be used is an FET; however,transistors of other types such as a bipolar transistor may be used, andthe selection of these may be determined by the design specificationdemanded in the power supply apparatus, the semiconductor manufacturingprocess to be used, and the like.

While the preferred embodiments of the present invention have beendescribed using specific terms, such description is for illustrativepurposes only, and it is to be understood that changes and variationsmay be made without departing from the spirit or scope of the appendedclaims.

1. A power supply apparatus for performing constant current driving of aload circuit, comprising: a constant current circuit which is disposedon a path for driving the load circuit; a voltage generating circuitstructured to output a driving voltage to the load circuit; and amonitoring circuit structured to monitor a voltage across the two endsof the constant current circuit; wherein the monitoring circuit includesa threshold voltage source structured to generate a threshold voltagethat follows the fluctuation of the voltage at which the constantcurrent circuit can operate stably, and wherein the monitoring circuitstructured to control the voltage generating circuit on the basis of aresult obtained by comparing the voltage across the two ends of theconstant current circuit and the threshold voltage generated by thethreshold voltage source, wherein the voltage generating circuit is acharge pump circuit which switch among a plurality of boosting ratios,and a boosting ratio of the charge pump circuit is switched on the basisof a result of voltage comparison of the monitoring circuit.
 2. Thepower supply apparatus according to claim 1, wherein a current generatedby the constant current circuit is adjustable according to a referencevoltage.
 3. The power supply apparatus according to claim 1, wherein theload circuit is a light emitting diode, and the constant current circuitis connected to a cathode terminal of the light emitting diode, and themonitoring circuit monitors the voltage at the cathode terminal of thelight emitting diode.
 4. The power supply apparatus according to claim1, wherein the voltage generating circuit is a switching regulatorcircuit, and a switching operation of the switching regulator circuit iscontrolled on the basis of a result of voltage comparison of themonitoring circuit.
 5. The power supply apparatus according to claim 4,further comprising: a control unit structured to receive an errorvoltage indicative of the result of voltage comparison of the monitoringcircuit, and wherein the control unit includes: an oscillator structuredto generate a periodic signal having a triangular wave shape; a voltagecomparator structured to compare the error voltage with the periodicsignal; and a driver structured to generate a signal for driving aswitching element of the switching regulator according to the output ofthe voltage comparator.
 6. A light emitting apparatus comprising: alight emitting diode, and a power supply apparatus according to claim 1for performing constant current driving of the light emitting diode. 7.An electronic equipment comprising: a liquid crystal panel, and a lightemitting apparatus according to claim 6 which is disposed as a backlight of the liquid crystal panel.
 8. A power supply apparatus forperforming constant current driving of a load circuit, comprising: aconstant current circuit which is disposed on a path for driving theload circuit; a voltage generating circuit structured to output adriving voltage to the load circuit; a monitoring circuit structured tomonitor a voltage across the two ends of the constant current circuit;and a control unit structured to receive an output signal indicative ofthe result of voltage comparison from the monitoring circuit, andcontrols the boosting ratio of the charge pump according to the outputsignal; wherein the monitoring circuit includes a threshold voltagesource structured to generate a threshold voltage that follows thefluctuation of the voltage at which the constant current circuit canoperate stably, and wherein the monitoring circuit structured to controlthe voltage generating circuit on the basis of a result obtained bycomparing the voltage across the two ends of the constant currentcircuit and the threshold voltage generated by the threshold voltagesource.
 9. A light emitting apparatus comprising: a light emittingdiode, and a power supply apparatus according to claim 8 for performingconstant current driving of the light emitting diode.
 10. An electronicequipment comprising: a liquid crystal panel, and a light emittingapparatus according to claim 9 which is disposed as a back light of theliquid crystal panel.
 11. A power supply apparatus for performingconstant current driving of a load circuit, comprising: a constantcurrent circuit which is disposed on a path for driving the loadcircuit; a voltage generating circuit structured to output a drivingvoltage to the load circuit; a monitoring circuit structured to monitora voltage across the two ends of the constant current circuit; and afirst oscillator structured to generate a first periodic signal having afrequency needed for switching control of the charge pump circuit;wherein the monitoring circuit includes a threshold voltage sourcestructured to generate a threshold voltage that follows the fluctuationof the voltage at which the constant current circuit can operate stably,and wherein the monitoring circuit structured to control the voltagegenerating circuit on the basis of a result obtained by comparing thevoltage across the two ends of the constant current circuit and thethreshold voltage generated by the threshold voltage source.
 12. Thepower supply apparatus according to claim 11,wherein the control circuitis structured to monitor the output signal of the monitoring circuit andis structured to raise the boosting ratio when a state in which theoutput signal is at a predetermined level continues for a predeterminedperiod of time, and wherein the power supply apparatus further includes:a second oscillator structured to generate a second periodic signalhaving a frequency needed for measuring the predetermined period. 13.The power supply apparatus according to claim 12, wherein the firstoscillator is in on state and the second oscillator is in off state whenthe boosting ratio is greater than 1, and the first oscillator is in offstate and the second oscillator is in on state when the boosting ratiois equal to
 1. 14. A light emitting apparatus comprising: a lightemitting diode, and a power supply apparatus according to claim 11 forperforming constant current driving of the light emitting diode.
 15. Anelectronic equipment comprising: a liquid crystal panel, and a lightemitting apparatus according to claim 14 which is disposed as a backlight of the liquid crystal panel.
 16. A power supply apparatus forperforming constant current driving of a load circuit, comprising: aconstant current circuit which is disposed on a path for driving theload circuit; a voltage generating circuit structured to output adriving voltage to the load circuit; and a monitoring circuit structuredto monitor a voltage across the two ends of the constant currentcircuit; wherein the monitoring circuit includes a threshold voltagesource structured to generate a threshold voltage that follows thefluctuation of the voltage at which the constant current circuit canoperate stably, and wherein the monitoring circuit structured to controlthe voltage generating circuit on the basis of a result obtained bycomparing the voltage across the two ends of the constant currentcircuit and the threshold voltage generated by the threshold voltagesource, and wherein the constant current circuit includes: a currentoutput terminal to which the load circuit to be driven is connected; anoperation amplifier having a first input terminal to which apredetermined reference voltage is applied; a first transistor having acontrol terminal to which an output voltage of the operation amplifieris applied and having one end connected to the current output terminal;a first resistor connected to the other end of the first transistor andhaving one end to which a predetermined fixed voltage is applied; and afeedback path structured to feed the electric potential of theconnection point of the first transistor and the first resistor back toa second input terminal of the operation amplifier, and the thresholdvoltage source includes: a constant current source structured to outputa predetermined constant current; a second transistor which is disposedin series on a path of the constant current that is output from theconstant current source; and a second resistor having one end to whichthe fixed voltage is applied and having the other end to which thesecond transistor is connected, and the constant current circuit isstructured to output the voltage of the connection point of the secondtransistor and the constant current source as the threshold voltage. 17.The power supply apparatus according to claim 16, wherein the secondtransistor and the first transistor, and the second resistor and thefirst resistor are formed respectively by pairing on a semiconductorintegrated circuit.
 18. A light emitting apparatus comprising: a lightemitting diode, and a power supply apparatus according to claim 16 forperforming constant current driving of the light emitting diode.
 19. Anelectronic equipment comprising: a liquid crystal panel, and a lightemitting apparatus according to claim 18 which is disposed as a backlight of the liquid crystal panel.